Fig. 6.29 Wavefonns of tlie push-pull isolated buck eouverler.

Conducting devices:

0 DT,

t>2

side conduttion losse.s, since at any given instant only one tiaiisistor is connected in series with the dc source V. The ability to operate with transistor duty cycles approaching unity also allows the turns ratio 11 to be minimized, reducing the transistor currents.

The push-pull configuration is prone to transformer saturation problems. Since it cannot be guaranteed that the forward voltage drops and conduction times of transistors and are exactly equal, small imbalances can cause the dc component of voltage applied to the transformer primary to be nonzero. In consequence, during every two switching periods there is a net increase in the magnitude of die magnetizing current. If this imbalance continues, then the magnetizing current can eventually become large enough to saturate the transformer,

Current-programmed control can be employed to mitigate the transformer saturation problems. Operation of the push-pull converter using only duty cycle control is not recommended.

Utilization of the transformer core material and secondary winding is similar to that for the full-bridge converter. The Ппх and magnetizing cmrent can be both positive and negative, and therefore the entire B~H loop can be used, if desired. Since the primary and .secondary windings are both center-tapped, their utilization is suboptimal.

Fig. 6.30 Derivation of the flyback converter: (a) buck-boost converter; (b) intluctor L is wound with two parallel wire, !; (c) inductor windiriES are isolated, leading to the flyback converter; (d) with a 1 ;n turns ratio aud positive output.

6.3.4 Flyback Converter

The flyback converter is based on the buck-boost converter. Its derivation is illustrated in Fig. 6.30. Figure 6.3(Xa) depicts the basic buck-boost converter, with the switch realized using a MOSFET and diode. In Fig. Й.30(b), the inductor winding is constructed using two wires, with a 1:1 turns ratio. The basic function of the inductor is unchanged, and the parallel windings are equivalent to a single winding constructed of larger wire. In Fig. 6.30(c), the connections between the two windings are broken. One winding is used while the transistor Qj conducts, while the other winding is used when diode conducts. The total current in the two windings is unchanged from the circuit of Fig. Й.30(b); however, the current is now distributed between the windings differently. The magnetic fields inside the inductor inboth cases

Fig. 6.31 hlybiick converter circuit: (a) with transformer equivalent circuit modci, fb) during subintcrvai 1, (c) during subitttervai 2.

j Transformer model \

о ; о

-I 2.

о

= 01 +

i Trimsformer model \

С dz. Л V

are identical. Although the two-winding magnetic device is represented using the same symbol as the transformer, a more descriptive name is two-winding inductor. This device is sometimes also called a flyback Iransformer. Unlike the ideal transformer, current does not flow simultaneously in both windings of the flyback transformer. Figure 6.3()(d) illustrates the usual configuration of the flyback converter. The MOSFET source is connected to the primary-side ground, simplifying the gate drive circuit. The transformer poliu-ity marks are reversed, to obtain a po,4itive output voltage. A 1: turns ratio is introduced; thi,4 allows better converter optimization.

The ilyback converter may be analyzed by insertion of the model of Fig. 6.16(b) in place of the flyback transformer. Tlie circuit of Fig. 6.31(a) is then obtained. The inagnetizing inductance L functions in the same manner as inductorL of the original buck-boost converter of Fig. fi.30(a). When transistor Q[ conducts, energy from the dc source V, is stored in L. When diode conducts, this stored energy is transferred to the load, with the inductor voltage and current scaled according to the l: turns ratio.

During subinterval 1, while transi.stor conducts, the converter circuit model reduces to Fig.